The present invention relates to a switching power supply circuit including a voltage resonant converter.
As types of a so-called soft switching power supply that employs a resonant converter, a current resonant type and a voltage resonant type have been widely known. Currently, half-bridge current resonant converters formed of a two-transistor switching element have been widely employed since they can easily be put into practical use.
However, since characteristics of high-breakdown-voltage switching elements are currently being improved for example, problems about breakdown voltage associated with putting voltage resonant converters into practical use are being cleared up. Furthermore, it is known that a single-ended voltage resonant converter formed of a one-transistor switching element is advantageous over a one-transistor current resonant forward converter with regard to input feedback noises and noise components of a DC output voltage line.
FIG. 19 illustrates one configuration example of a switching power supply circuit including a single-ended voltage resonant converter, disclosed in Japanese Patent Laid-Open No. 2000-134925.
In the switching power supply circuit of FIG. 19, a voltage from a commercial alternating-current power supply AC is rectified and smoothed by a rectifying and smoothing circuit formed of a bridge rectifier circuit Di and a smoothing capacitor Ci, to thereby generate a rectified and smoothed voltage Ei as the voltage across the smoothing capacitor Ci.
The lines from the commercial power supply AC are provided with a noise filter that includes a pair of common mode choke coils CMC and two across-line capacitors CL, and removes common mode noises.
The rectified and smoothed voltage Ei is input to the voltage resonant converter as a DC input voltage. The voltage resonant converter has a single-ended configuration including a one-transistor switching element Q1 as described above. The voltage resonant converter in this circuit is separately excited. Specifically, the switching element Q1 formed of a MOS-FET is switch-driven by an oscillation and drive circuit 2.
A body diode DD of the MOS-FET is connected in parallel to the switching element Q1. In addition, a primary-side parallel resonant capacitor Cr is connected in parallel to the channel between the drain and source of the switching element Q1.
The primary-side parallel resonant capacitor Cr and the leakage inductance L1 of a primary winding N1 in an isolation converter transformer PIT form a primary-side parallel resonant circuit (voltage resonant circuit). This primary-side parallel resonant circuit offers voltage resonant operation as the switching operation of the switching element Q1.
In order to switch-drive the switching element Q1, the oscillation and drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1. Thus, the switching element Q1 implements switching operation with the switching frequency dependent upon the cycle of the drive signal.
The isolation converter transformer PIT transmits switching outputs from the switching element Q1 to the secondary side.
The isolation converter transformer PIT is constructed of an EE-shaped core that is formed by combining E-shaped cores composed of a ferrite material for example. Furthermore, the primary winding N1 and a secondary winding N2 are wound around the center magnetic leg of the EE-shaped core, with the winding part being divided into the primary side and secondary side.
In addition, a gap with a length of about 1.0 mm is provided in the center leg of the EE-shaped core in the isolation converter transformer PIT, so that a coupling coefficient k of about 0.80 to 0.85 is obtained between the primary side and the secondary side when the coupling coefficient k has such a value, the coupling degree between the primary and secondary sides may be regarded as loose coupling, and thus it is difficult to obtain the saturation state. The value of the coupling coefficient k is a factor in setting the leakage inductance (L1).
One end of the primary winding N1 in the isolation converter transformer PIT is interposed between the switching element Q1 and the positive electrode of the smoothing capacitor Ci. Thus, the transmission of switching outputs from the switching element Q1 is allowed. In the secondary winding N2 of the isolation converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated.
On the secondary side, a secondary-side series resonant capacitor C2 is connected in series to one end of the secondary winding N2, and therefore the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary-side series resonant capacitor C2 form a secondary-side series resonant circuit (current resonant circuit).
Furthermore, connected to this secondary-side series resonant circuit are rectifier diodes Do1 and Do2 and a smoothing capacitor Co, to thereby form a voltage-doubler half-wave rectifier circuit. This voltage-doubler half-wave rectifier circuit generates, as the voltage across the smoothing capacitor Co, a secondary-side DC output voltage Eo with the level twice that of an alternating voltage V2 induced in the secondary winding N2. The secondary-side DC output voltage Eo is supplied to a load, and is input to a control circuit 1 as a detected voltage for constant-voltage control.
The control circuit 1 detects the level of the secondary-side DC output voltage Eo input as a detected voltage, and then inputs the obtained detection output to the oscillation and drive circuit 2.
According to the level of the secondary-side DC output voltage Eo indicated by the detection output, the oscillation and drive circuit 2 controls the switching operation of the switching element Q1 so that the secondary-side DC output voltage Eo is kept constant at a certain level. That is, the oscillation and drive circuit 2 generates and outputs a drive signal for achieving intended switching operation. Thus, stabilization control of the secondary-side DC output voltage Eo is achieved.
FIGS. 20A to 20C and 21 show results of experiments on the power supply circuit in FIG. 19. For the experiments, the power supply circuit of FIG. 19 was designed to include major parts with the following characteristics.
The core of the isolation converter transformer PIT employed an EER-35 core, and a gap in the center leg thereof was designed to have a gap length of 1 mm. The numbers of turns of the primary winding N1 and the secondary winding N2 were set to 39 T and 23 T, respectively. The induction voltage level per one turn (T) in the secondary winding N2 was set to 3 V/T. The coupling coefficient k of the isolation converter transformer PIT was set to 0.81.
The capacitance of the primary-side parallel resonant capacitor Cr was set to 3900 pF. The capacitance of the secondary-side series resonant capacitor C2 was set to 0.1 μF. Accordingly, the resonant frequency fo1 of the primary-side parallel resonant circuit was set to 230 kHz, and the resonant frequency fo2 of the secondary-side series resonant circuit was set to 82 kHz. Therefore, the relative relationship between the resonant frequencies fo1 and fo2 can be represented as fo1≈2.8×fo2.
The rated level of the secondary-side DC output voltage Eo was 135 V. The allowable load power range was from the maximum load power Pomax of 200 W to the minimum load power Pomin of 0 W.
FIGS. 20A to 20C are waveform diagrams showing the operation of the major parts in the power supply circuit in FIG. 19, with reflecting the corresponding switching cycle of the switching element Q1. FIG. 20A shows a voltage V1, a switching current IQ1, a primary winding current I1, a secondary winding current I2, and secondary-side rectified currents ID1 and ID2, when the load power is the maximum load power Pomax of 200 W. FIG. 20B shows the voltage V1, the switching current IQ1, the primary winding current I1, and the secondary winding current I2, when the load power is intermediate load power Po of 120 W. FIG. 20C shows the voltage V1 and the switching current IQ1 when the load power is the minimum load power Pomin of 0 W.
The voltage V1 is the voltage obtained across the switching element Q1, and has a waveform like those in FIGS. 20A to 20C. Specifically, the voltage level is at 0 level during the period TON when the switching element Q1 is in the on-state, while a sinusoidal resonant pulse is obtained during the period TOFF when it is in the off-state. This resonant pulse waveform of the voltage V1 indicates that the operation of the primary-side switching converter is voltage resonant operation.
The switching current IQ1 is the current flowing through the switching element Q1 (and the body diode DD). The switching current IQ1 flows with the illustrated waveforms during the period TON, while it is at 0 level during the period TOFF.
The primary winding current I1 flowing through the primary winding N1 is the current resulting from the synthesis between the current flowing as the switching current IQ1 during the period TON and the current flowing to the primary-side parallel resonant capacitor Cr during the period TOFF.
The rectified currents ID1 and ID2, illustrated only in FIG. 20A, flowing through the rectifier diodes Do1 and Do2 as the operation of the secondary-side rectifier circuit have sinusoidal waveforms like the illustrated ones. The waveform of the rectified current ID1 indicates the resonant operation of the secondary-side series resonant circuit more dominantly than the waveform of the rectified current ID2.
The secondary winding current I2 flowing through the secondary winding N2 has a waveform resulting from the synthesis between the waveforms of the rectified currents ID1 and ID2.
FIG. 21 shows, as functions of the load, the switching frequency fs, the lengths of ON and OFF periods TON and TOFF of the switching element Q1, and the AC to DC power conversion efficiency (ηAC→DC) of the power supply circuit shown in FIG. 19.
Referring initially to the AC to DC power conversion efficiency (ηAC→DC), it is apparent that high efficiencies of 90% or more are achieved for a wide range of the load power Po from 50 W to 200 W. The inventor of the present application has previously confirmed, based on experiments, that such a characteristic is obtained when a single-ended voltage resonant converter is combined with a secondary-side series resonant circuit.
In addition, the switching frequency fs, the period TON, and the period TOFF in FIG. 21 indicate the switching operation of the power supply circuit in FIG. 19 as the characteristic of constant-voltage control against load variation. In the power supply circuit, the switching frequency fs is almost constant against the load variation. In contrast, the periods TON and TOFF show linear changes having opposite tendencies as shown in FIG. 21. These characteristics show that the switching operation is controlled against the variation of the secondary-side DC output voltage Eo such that the time ratio between the ON and OFF periods is changed with the switching frequency (switching cycle) being kept almost constant. This control can be regarded as pulse width modulation (PWM) control, in which the lengths of the ON and OFF periods within one switching cycle are changed. This PWM control allows the power supply circuit in FIG. 19 to stabilize the secondary-side DC output voltage Eo.
FIG. 22 schematically shows the constant-voltage control characteristic of the power supply circuit shown in FIG. 19, based on the relationship between the switching frequency fs (kHz) and the secondary-side DC output voltage Eo.
The power supply circuit shown in FIG. 19 includes a primary-side parallel resonant circuit and a secondary-side series resonant circuit, and therefore has two resonant impedance characteristics in a complex manner: the resonant impedance characteristic corresponding to the resonant frequency fo1 of the primary-side parallel resonant circuit, and that corresponding to the resonant frequency fo2 of the secondary-side series resonant circuit. Since the power supply circuit in FIG. 19 has the frequency relationship fo1≈2.8×fo2, the secondary-side series resonant frequency fo2 is lower than the primary-side parallel resonant frequency fo1 also as shown in FIG. 22.
The characteristic curves in FIG. 22 show constant-voltage control characteristics in association with control of the switching frequency fs, assumed based on these resonant frequencies and under the condition of a certain constant AC input voltage VAC. Specifically, Characteristic curves A and B indicate the constant-voltage control characteristics obtained when the load power is the maximum load power Pomax and when it is the minimum load power Pomin, respectively, based on the resonant impedance corresponding to the resonant frequency fo1 of the primary-side parallel resonant circuit. Characteristic curves C and D indicate the constant-voltage control characteristics obtained when the load power is the maximum load power Pomax and when it is the minimum load power Pomin, respectively, based on the resonant impedance corresponding to the resonant frequency fo2 of the secondary-side series resonant circuit. When, under the characteristics in FIG. 22, constant-voltage control is intended so that the output voltage is kept at the voltage tg that is the rated level of the secondary-side DC output voltage Eo, the variation range of the switching frequency fs required for the constant-voltage control (requisite control range) can be expressed by the area indicated by Δfs.
The requisite control range Δfs shown in FIG. 22 is from the frequency offering the voltage level tg on Characteristic curve C, corresponding to the resonant frequency fo2 of the secondary-side series resonant circuit and the maximum load power Pomax, to that on Characteristic curve B, corresponding to the resonant frequency fo1 of the primary-side parallel resonant circuit and the minimum load power Pomin. The range Δfs includes the frequency offering the voltage level tg on Characteristic curve D, corresponding to the resonant frequency fo2 of the secondary-side series resonant circuit and the minimum load power Pomin, and that on Characteristic curve A, corresponding to the resonant frequency fo1 of the primary-side parallel resonant circuit and the maximum load power Pomax.
Therefore, as constant-voltage control operation, the power supply circuit in FIG. 19 implements switching drive control based on PWM control in which the time ratio of the periods TON/TOFF in one switching cycle is changed with the switching frequency fs being kept almost constant. The implementation of the PWM control is indicated also by FIGS. 20A to 20C, in which the widths of the periods TOFF and TON change depending on the load power while the length of one switching cycle (TOFF+TON) is almost constant when the maximum load power Pomax of 200 w, load power Po of 120 w and the minimum load power Pomin of 0 w.
Such operation is due to the resonant impedance characteristic of the power supply circuit against load variation. Specifically, carried out under the narrow switching frequency range (Δfs) is transition between the state where the resonant impedance corresponding to the resonant frequency fo1 of the primary-side parallel resonant circuit (capacitive impedance) is dominant, and the state where the resonant impedance corresponding to the resonant frequency fo2 of the secondary-side series resonant circuit (inductive impedance) is dominant.
The power supply circuit in FIG. 19 involves the following problems.
Referring to the aforedescribed waveform diagrams of FIGS. 20A to 20C, the switching current IQ1 when the load power is the maximum load power Pomax, shown in FIG. 20A, operates as follows. Specifically, the switching current IQ1 is at 0 level until the end of the period TOFF, which is the turn-on timing of the switching element Q1. When the period TON starts, initially a current of the negative polarity flows through the body diode DD, and then the polarity is inverted and the switching current IQ1 flows between the drain and source of the switching element Q1. This operation indicates the state where zero voltage switching. (ZVS) is adequately carried out.
In contrast, the switching current IQ1 when the load power is the intermediate load power Po of 120 W, shown in FIG. 20B, shows a waveform in which a noise current flows at timing immediately before the end of the period TOFF, which is the turn-on timing of the switching element Q1. This waveform indicates abnormal operation in which ZVS is not implemented adequately.
That is, it is known that a voltage resonant converter including a secondary-side series resonant circuit as shown in FIG. 19 involves abnormal operation in which ZVS is not implemented adequately when the load is an intermediate load. It has been confirmed that, in an actual power supply circuit of FIG. 19, such abnormal operation arises in the load variation range indicated by the area A in FIG. 21 for example.
A voltage resonant converter including a secondary-side series resonant circuit originally has a tendency to have a characteristic of keeping high efficiencies favorably against load variation as described above. However, as shown with the switching current IQ1 of FIG. 20B, a corresponding peak current flows at the turn-on timing of the switching element Q1. This noise current causes an increase of switching loss, which is a factor in a decrease of the power conversion efficiency.
In addition, the occurrence of such abnormal operation anyway yields an offset of the phase-gain characteristic of the constant-voltage control circuitry for example, which leads to switching operation in an abnormal oscillation state. Therefore, currently there is strong recognition of actual difficulty in putting the voltage resonant converter into practical use.